Posts related to power products:
Simulating Power Supply Noise

Designs ranging from transceiver chips for smart phones to radar assemblies for fighter jets require power supply noise immunity testing, which is often referred to as power supply rejection ratio (PSRR) testing. Characterizing the design’s ability to reject or attenuate supply noise is critical because the noise can manifest itself as bit errors in the case of transceiver chips and inaccurate target data in the case of radar assemblies. In this post, we will discuss the challenges involved in simulating power supply noise signals for PSRR testing and look at two economical methods for doing it.

Challenges in simulating power supply noise
When you test PSRR, one challenge is modulating a quantitative noise signal onto a DC power supply level. This task is difficult because the power supply inputs typically present extremely low input impedance to any AC signal because of the high input parallel capacitance. While high-performance power supplies do have built-in arbitrary waveform capability, the output bandwidth of power supply waveforms is typically limited to less than 30 KHz. This limitation only allows you to simulate low frequency noise sources, such as power line ripple. It falls well short of the more critical noise source frequency ranges, such as power supply switching noise which ranges from 50 KHz to greater than 20 MHz.

To put the first challenge in perspective, let’s consider an example transceiver chip for a smart phone. The power supply input most likely has a parallel 1-uF bypass ceramic capacitor as well as a large 10-uF electrolytic capacitor to serve as an energy reservoir for the sharp transmit transients. That means a 1-MHz sinewave modulated onto the power supply level of the RF power amplifier would, in theory, only see a load of less than 20 mOhms. In practice, the equivalent series resistance (ESR), equivalent series inductance (ESL), and other parasitic impedances boosts the load impedance up to more than 50 mOhms, but that still presents low input impedance to any AC signal content.

Let's look at two solutions for simulating power supply noise that use (for the most part) general-purpose test equipment and simple components that you can find around the office or lab.

The op amp/MOSFET solution (OMS)
The OMS solution can be seen in the below figure. The solution requires two power supplies, a function/arbitrary waveform generator (FG/AWG), an op amp, and a high-power N-type MOSFET. In Figure 1, C1, C2, and the resistance value labeled “Load” make up a circuit representation of the device under test’s power supply input.

The OMS solution uses the MOSFET as a variable resistor between the power supply and the DUT (load). By varying the MOSFET’s drain to source resistance we can modulate the power supply's DC level with the desired noise signal. The op amp is used to maintain the desired voltage level at the DUT. With the op amp's inverting input connected to the same node as the input to the load, the op amp will drive the MOSFET's gate such that the amplitude level at its inverting input is the same as the amplitude at its non-inverting input.

The FG/AWG’s output creates the amplitude level and noise signal at the DUT's input. The power supply output voltage should be set comfortably higher than the desired amplitude plus any added noise signal amplitude to ensure the MOSFET stays properly biased. This OMS circuit allows you to accurately generate noise signals with bandwidths up to about 500 KHz. The bandwidth can vary depending on the DUT's input impedance and how well the OMS circuit was constructed. For best results, keep wiring and leads as short as possible to prevent oscillations.

As an example, the OMS circuit was implemented using an Agilent 33521A function/arbitrary waveform generator, N6762A power supply, E3630A dual-output power supply (to power the op amp), an IRFP150N N-type MOSFET, and an NE5534A op amp. The below scope capture shows the example OMS implementation generating a 200-mVpp 500-KHz sine wave onto a 5-VDC level. The same DUT load shown in the above diagram was used for this example, the 100-uF capacitor is electrolytic and the 1-uF capacitor is ceramic.

When you implement the OMS, you will need to use a high-performance power supply because you need a supply with a fast transient response and a stable output to ensure the supply can handle the rapidly varying load levels caused by the noise signal. Also, in the OMS the DC level value is limited to the max voltage of the FG/AWG, which today is typically 10 V. To overcome this limitation, you could add a voltage amplifier between the op amp and the FG/AWG.

Amplified modulation solution (AMS)
The AMS does not have the voltage or bandwidth limitations that the OMS has, but it does have a higher price tag. The AMS is shown in the below figure. The AMS consists of a power supply, FG/AWG or signal generator, wide-band or RF power amp and coupling capacitor section.

The FG/AWG and RF amp are used to modulate the DC level from the power supply with the desired noise signal. The coupling capacitor section blocks the DC level from the RF amp's input and provides a low-impedance path for the AC output of the power amp. The power amplifier is needed to boost the noise signal power to deal with the low impedance to AC that the DUT's input presents. You'll want to make the coupling capacitor section's impedance as low as possible, so you may want to use multiple capacitor values and types in parallel to reduce the effects of ESR and ESL. The exact circuit setup, the DUT's true input impedance, and the bandwidth of the noise signal will all determine how many watts the RF amp should be. A good place to start is 50 W, and you can go up from there. Note that the RF amp's output impedance should be as low as possible, which is typically going to be 50 ohms. With the DUT's impedance being so low, the RF amp will see a large amount of reflected power at its input, so when you choose an RF amplifier, be sure to select one that can handle a short-circuit condition.

As an example, the AMS circuit was implemented using an Agilent 33521A function/arbitrary waveform generator, an N6762A power supply, an E&I 1040L RF power amplifier, and 10-uF and 0.1-uF capacitors in parallel (for the coupling capacitor section). A 1040L RF amp provided up to 400 W of output power, which was more than necessary, but I used what was available to me in the lab. Using the same load used in the OMS example, the AMS was used to create a single tone noise signal. The below scope capture shows a 1-Vpp 5-MHz sinewave was modulated onto a 10-VDC level.

Test and measurement power supplies typically have a high capacitance value at their output. In the AMS, the power supply is in parallel with the load in regards to the RF amp’s output. That means the noise signal actually has two low-impedance paths to take, the load and the power supply output. It is hard to know the power supply’s AC power handling characteristic across its input capacitance since it is not specified by the power supply manufacturer, so you may want to contact the manufacturer to avoid costly damage.

One safety precaution you can take is adding blocking impedance between the power supply output and the coupling capacitor section. The blocking impedance can either be an inductor or a resistor. Its purpose is to protect the power supply from being damaged by the noise signal by adding higher series impedance before the power supply output. The inductor is ideal because it acts like a short to the DCV from the power supply, but presents high impedance to the AC noise signal. However, the inductor can cause large oscillations on the DC level if the DUT has a very dynamic current.

Using a resistor avoids the oscillations problem. Be sure to use the power supply’s sense leads to regulate the output voltage at the load to compensate for the voltage drop across the resistor. Also be sure to use a low-value resistor no higher than 1 Ohm. This is to ensure a low voltage drop across the resistor so you do not damage the sense lead circuit on the power supply.

PSRR testing is required on a wide range of devices today from smart phone chips to electronic assemblies for aircraft. Testing PSRR includes generating high-frequency noise signals on power supply levels. This task is challenging due to the high capacitance present at most power supply inputs. In this article we looked at two low-cost solutions for modulating noise signals on power supply levels. If you have any questions on the post send me an email and if you have anything to add use the "Comments" section below.

Testing DC to DC Converters with a Single Instrument

This post features two videos by a colleague of mine who presents a single instrument solution for testing DC to DC converters targeted for handheld or portable devices where power optimization is critical. Typically when testing a DC to DC converter you need a long list of test equipment including a power supply, eload, multi-channel scope or digitizer, current shunt or probe, and a waveform generator to vary the output of the power supply or eload. Agilent's N6705B DC Power Analyzer provides a single instrument solution. The N6705B is a modular power supply with up to four power outputs, voltage and current digitizers, and power waveform capabilities. There are over 30 power modules to choose from. The SMU modules shown in the video can source and sink power so one is connected to the input of the DC to DC converter and the other sits on the output as a load.

The first video provides an overview and example measurements for using the N6705B for testing DC to DC converters. The second video goes into more detail on a DC to DC converter efficiency measurement and shows an example.

How to get Your Electronic Load Down to Zero Volts

Typically a DC electronic load (eload) cannot go down to zero volts (they cannot sink current at zero volts) and their current handling and other performance specs begin to derate below 3 volts. This can be a big problem when you need to test low voltage power sources, like photovoltaic panels and power supplies for FPGAs. In this post we will look at why eloads begin to derate at low voltages and cannot go down to zero volts and look at simple configuration that will let you test power sources with an eload down to zero volts.

Most eloads have limited operation below three volts. To understand these limitations lets look at a simplified diagram of a typical eload in the below figure.

The FET acts like a shunt resistor across the power supply under test. As the transistor turns on harder it draws more current from the power supply under test. The power generated by the power supply is therefore dissipated in the load transistor. As long as the power supply output voltage is sufficient to bias the load transistor everything works fine. However, if the power supply voltage across Vds is low, about 3 volts or less, the load transistor can no longer regulate the current. At the point Vds minimum is reached, the load transistor is turned on to full saturation and the load it presents to the Power Supply under test is simply its saturation resistance, Rdson. As an example the below figure shows the operating curve for Agilent's N3304A Electronic Load.

Notice in the figure, below 3 volts the load can be used at reduced current but it will have poor dynamic (transient) response due to the fact that the transistor is in saturation.

A solution to the low voltage dilemma is to insert an auxiliary boost power supply in series with the electronic load and the power supply under test as shown in the below figure. The auxiliary supply "boosts" the supply under test's floating low such that it always appears to be at a voltage potential at or above the voltage potential of the auxiliary supply to the eload.

Adding the sense lines connected across the supply under test compensates for the boost supply voltage and the cabling in the measurement. Without the sense lines you would just have to subtract the boost supply voltage from the eload's voltage measurement. The boost supply can be a low-cost fixed output 3V to 5V power supply with current rating at least as high as the maximum peak load current needed. While this configuration will compensate for the load minimum voltage requirement and voltage drop in the power leads it has some disadvantages explained next.

Three factors must be considered when using the above configuration:
  1. Any voltage inaccuracies (when sense is not used) or current noise introduced by the boost supply will affect the measurement accuracy of the eload. 
  2. The eload must have a high enough power rating to dissipate the power from the supply under test and the boost supply.
  3. There is a possibility that the boost supply could reverse bias the power supply under test as the voltage across the load decreases. This can occur, for example, when the power supply under test can no longer maintain its output voltage because it is in overcurrent protection mode.
Fortunatly high performance eloads like Agilent's N3300 eload series employ detection circuits to prevent the third factor from occurring.

In this post we looked at why eloads cannot operate at full current and performance levels at low voltages. We also covered a configuration using a boost supply that allows you to overcome an eload's low voltage limitations and to test a power source all the way down to zero volts. If you have anything to add to this post use the comments section below.

On Demand Webcast: Simulating Power Transients and Noise

Recently I delivered a webcast entitled "Simulating Power Transients and Noise." The webcast was recorded and is now available on demand. Below is an abstract on webcast, followed by the link to access it. The content is focused on the aerospace defense industry, but is applicable to various industries such as wireless and automotive.

DC powered Shipboard, ground vehicle, aircraft, and space bound electronic circuits and assemblies employ a robust input power supply design to withstand non-ideal input power states. These non-ideal states arise from operating conditions such as large load changes on the main power source, harsh operating environments, and sudden switches from one main power source to another. These non-ideal states take the form of transients on the DC supply level or as noise signals coupled onto the DC supply level. During design these non-ideal power supply states must be simulated to test the input power supply design’s robustness against these conditions.

In this webcast we will look how easy it is to create arbitrary power waveforms on modern programmable power supplies for simulating supply transients and noise. We will also look at the technology hurdles that limit the arbitrary waveform bandwidth on modern supplies. Finally we will look at low cost methods for simulating fast power supply transients and high frequency noise beyond the bandwidth limitations of modern power supplies.

Access Webcast: Simulating Power Transients and Noise

Configuring a Split Rail Power Supply

In this post we will look at creating a split rail power supply that provides both a positive and negative voltage output. This is a trivial task if you have a multiple output power supply that has a negative output lead. This task becomes more challenging when all you have at your bench is single quadrant power supplies that can only output positive voltage and positive current.

This challenge of creating a split rail supply can be overcome by combining two single quadrant supplies in series. If we call the supply that creates the negative supply voltage V1 and the supply that creates the positive voltage V2, the high lead of V1 will be connected to the low lead of V2. This forms the common or "rail" node of the supplies, which can be tied to ground or to the common of the DUT. The simple split rail power supply setup can be seen in the figure below.

Note that the high lead of V2 is then connected to the positive supply input of the DUT and the low lead of V1 is connected to the negative supply input of the DUT. Lets look at an example where we want to create a split rail supply with a 15 V output and a -15 V output. both outputs were connected to a DMM and the resulting measurements can be seen in the figure below.

Note that the common or low leads on both DMMs are at the same potential. When implementing this setup be sure that the low lead of the negative voltage supply (V1 in the first figure) is not tied to ground. Additional supply outputs, either positive or negative, can be easily added. This is done by either tying the additional power supply low lead, in the case of a positive output, or high lead, in the case of a negative output to the rail node.

In this post we looked at how to configure two single quadrant power supplies to power a DUT that requires a split rail power supply. If you have questions please shot me an email and if you have any personal insights to add to this post please use the comments section below.

ECU Fuel Injection Loop Calibration

In this post we will look at how a precision DC source or an SMU can be used to calibrate the fuel injection loop in a automobile or other engine powered equipment. The fuel injection flow to an engine is controlled by a solenoid that is adjusted either more open or more closed by a driver output from the engine control unit (ECU). Fuel injection control has always been an important part or a vehicle design, but recently the precision it is tested to has become even more critical because it is closely tied to a vehicle's fuel efficiency. Tight control over fuel injection means at any given moment the engine is getting the right mixture of fuel and air to ensure maximum efficiency.

Below is a basic circuit diagram of a fuel injection control circuit. The inductor at the top represents the fuel injector solenoid. The amount of current flowing through the solenoid determines how open or closed it is, which in turn controls the flow of fuel to the engine. The amount of current flowing through the solenoid is controlled by the MOSFET, which is pulse width modulated by a control circuit from the ECU. The rate of the PWM is determined using the control signal from the accelerator and the output of the current sense circuit. The test we will look at is used to calibrate the control loop to the output of the current sense circuit at various current levels.

There are two ways the calibration factors could be determined. The first is by isolating the circuit from  Vbatt and the solenoid and just forcing a known amount of current through the circuit using a current source. Below is a an example of how the test was done by an engineer at an automotive company using the Agilent N6762A Precision DC Source.

The DC source is used in constant current mode so it acts like a current source. Precision DC sources have high accuracy current sourcing capabilities which is ideal for this application. To perform the test iterate the output current value of the DC source through the full range of the solenoid, open to closed. Then use the measured values to build a look up table or create a curve fitted equation. Instead of a precision DC source a source measure unit (SMU) could also be used. An SMU will typically deliver slightly better current accuracy, but at a higher price. 

The second way to calibrate the fuel injection control is by operating the whole circuit in normal operation with Vbat and the solenoid or at least something that simulates Vbatt and the solenoid. You then use an SMU as a zero ohm shunt in series with the current path. To use an SMU as a zero ohm shunt you simply insert it into the circuit set it for 0V so its output is regulated at 0V. That way there is no voltage dropped across it so it doesn't effect the circuit. The circuit current from Vbatt will then flow through the SMU (it is sinking the current). The SMU would then accurately measure the current flowing through it. Using an SMU as a zero ohm shunt is much better then using a real current shunt, because a real shunt will add a non-zero amount of resistance to the current path. For this type of setup you want to make sure the SMU has a fast transient response spec to ensure proper regulation at 0 volts in the face of dynamic current changes.

In this post we looked at two ways to calibrate the fuel injection control loop in an automobile using either a precision DC source or an SMU. If you have any questions on this post please email me and if you have any insights to add just use the "Comments" section below.

Fine Tuning the FPGA Power-on Process

Powering on an FPGA can be careful balancing act of proper sequencing and slew rate timing on multiple power supply inputs. Some of the reasons for this balancing act is to ensure the power-on-reset circuit is not tripped, to ensure proper operation of the PLL, and to minimize in-rush current which is especially important in low power applications. In this post we will look at how modern multiple output power supplies can be used for fine tuning the FPGA turn-on process with an emphasis on minimizing in-rush current.

Modern multiple output power supplies have three features that make them a valuable tool for tuning the FPGA turn-on process, they are:
  • Adjustable output sequencing feature that allows the users to setup the turn-on time for each of the power supply's outputs.
  • Slew rate adjustment to specify the rise time at turn-on for each power supply output.
  • Output current and voltage digitizers for capturing the in-rush current, spotting voltage sags, etc.
The modern multiple output power supply comes into the FPGA circuit design process before implementing the power distribution system for the circuit. Use the modern multiple output power supply to tune the sequencing turn-on timing and slew rate to find the ideal turn-on conditions for the FPGA circuit. While tuning the turn on timing, the power supply measurement digitizers are used to measure the resulting voltage and the resulting in-rush current at turn-on. After the tuning process, the power distribution system can be designed and setup to achieve the ideal sequence and slew rate timing established earlier with the modern multiple output power supply.

Let's look at a quick example using a low power FPGA circuit that is going into a portable battery powered device. The engineer designing the power distribution system for the FPGA circuit wants to keep power usage to a minimum, which includes tuning the power turn-on conditions such that in-rush current is kept to a minimum amount while ensuring proper turn-on of the circuit. To tune the power turn-on conditions the engineer is using Agilent's N6705B DC Power Analyzer as the modern multiple output power supply. The N6705B is a modular power supply that supports up to 4 outputs. The N6705B has output sequencing, slew rate control, and output measurement digitizers along with a scope like display to analyze the digitized measurements. 

The FPGA circuit under test has four supply inputs. The engineer sets each of the N6705B outputs according to the specified ranges of the FPGA. A test run was done to verify the sequencing and slew rate of the power supply outputs. The result was captured on the N6705B's display and it can be seen in the below figure.

V1, V2, and V3 were all used for powering the FPGA. V4 was used for other components in the circuit. V1 is the FPGA's core supply and that is where we want to measure the in-rush current. Using the timing parameters shown in the above figure, the in-rush current on the core supply was captured and is shown in the figure below. 

As you can see in the figure the in-rush current consists of two spikes. The first is just over 1 ms with a flat top and the second is a higher current shorter duration spike. After the in-rush current, the low power FPGA's standby static current settles to about 15 mA. The engineer then did a number of tuning iterations of the same test. At each iteration the sequencing or the slew rate time was adjusted with the goal of minimizing the in-rush current. After tuning the engineer was able to get the in-rush current down to essentially zero (see figure below) while still maintaining a proper turn-on of the circuit.

In this post we looked at how a modern power supply with features such as output sequencing, slew rate control, and measurement digitizers can be used to tune the power input turn-on characteristics of an FPGA or any other embedded circuit for designing the power distribution. This is valuable when trying to ensure proper turn-on every time and to minimize in-rush current for power optimization. Below is a link to get more information on Agilent's N6705B DC Power Analyzer and also a link to a good TI app note that discusses properly powering on an FPGA. If you have any questions just email me and if you have any comments use the "Comments" section below.

Building an Electronic Load with General Purpose Test Equipment

In this post we will look at how to build a low cost DC electronic load (eload) using general purpose test equipment found on every engineer's test bench and some simple circuit components. The eload we build can be used for testing and validating power supplies and other power sources.

The below figure shows the eload circuit. The only piece missing from the figure is the measurement instrument(s) which are used to measure the voltage across the eload and the current through the eload. To measure voltage or current a scope, DMM, or digitizer could be used. The measurement points for the voltage and current measurements are mark in the figure as "Vmeas" and "Imeas" respectively.

Circuit Diagram of the eload
 The following list explains each part of the eload circuit and its function:

  • The device under test (DUT) is the DC power source we are testing with our eload circuit.
  • The N-Type MOSFET serves as a variable resistor and is the main component making up the load that the DUT is connected to. Its resistance is dynamically controlled by the op amp. The MOSFET will be dissipating the majority of the DUT's output power so power handling is a critical spec when choosing a MOSFET. You also want to ensure the MOSFET has proper heat sinking.
  • R1 serves as a sense resistor. The voltage drop across R1 is used by the op amp to control the current output of the DUT. R1 is also used to measure the output current from the DUT.
  • The op amp dynamically controls the resistance of the eload. It works by taking a control voltage level into its non-inverting input. It will then drive the gate of the MOSFET so that the same voltage value is at its inverting input, which is connected to the R1 node. This means the op amp allows us to control the voltage drop across R1 regardless if there is a change in the DUT's output. Since we know the value of R1 we can control the amount of current through R1. Since R1 is in series with the load, this means we can control the output current of the DUT.
  • The function / arbitrary waveform generator (FG / AWG) provides the control voltage to the op amp, which means it determines the output current of the DUT. The FG / AWG can be used as a simple DC bias to create a static output power from the DUT or a waveform could be used to create a dynamic load change (we look at an example of this later).
  • A basic bench top power supply can be used to power the op amp. You can use a bipolar supply to provide both the positive and negative supply voltages for the op amp.
  • To measure the voltage across the eload or the current through it a scope, DMM, or digitizer can be used. To measure both simultaneously you will need two channels. The voltage is measured from the drain of the MOSFET to ground. The current is measured by measuring the voltage drop across R1 and using the value of R1 and Ohm's Law to calculate current. The DMM is only good for static measurements. The scope or digitizer should be used when trying to capture dynamic voltage and current measurements.
Now that we know how the eload circuit works lets look at an example implementation of it measuring the transient response on a 10 V power supply. For this example we want to measure the supply's transient response time (within 3% of 10 V) to a load change from 0.5 A to 2.5 A. The following instruments and components were used for the implementation of the eload circuit:
  • 33521A function / arbitrary waveform generator
  • E3630A DC power supply
  • MSO7054A scope (4 analog channels)
  • NE5534A op amp. Chosen for high output bandwidth and low noise output.
  • IRFP150N power MOSFET. Chosen for its high power handling and low "on" resistance. Used with a metal heat sink.
  • 1 ohm 30 W power resistor
To create a sharp repeating load transient that ranged from 0.5 A to 2.5 A, the 33521A was setup to output a pulse signal that ranged from 0.5 V (0.5 V * 1 Ohm = 0.5 A) to 2.5 V (2.5 V * 1 Ohm = 2.5 A). Below is a screen capture from the scope showing the voltage across the eload (yellow) which is the power supply's output voltage and the voltage drop across the eload's sense resistor (green) which is the power supply's output current since the sense resistor is 1 Ohm. The blue trace is the pulsed output of the 33521A FG / AWG.

Notice from the green trace the load toggles between drawing 0.5 A and 2.5 A. This is equivalent to the eload toggling between 20 Ohms and 4 Ohms. Notice the positive and negative spikes on the yellow voltage trace, they are power supply transients caused by the sharp load changes, this is what we want to measure.

In the below figure, using the scope cursors, the power supply's transient response time was measured to within 3% of 10 V.

As you can see in the above figure the transient response time of the supply was < 1.4 us. We can also see the max deviation of the supply's voltage level due to the sudden load change, which is about 1.25 V or 12.5% of the nominal voltage level.

One thing you have to watch out for when building the eload circuit is oscillations. To prevent oscillations be sure to keep all wiring, connections, and traces as short as possible. Use a large ground plane and be careful to avoid ground loops. You can also add coupling capacitors as needed. 

In this post we looked at how to build a simple bench top DC electronic load for testing power supplies and other DC power sources. The eload design employs some general purpose test equipment and basic circuit components. The best part about the eload design that was presented in this post is it is essentially free since it uses common benchtop instruments and basic components that can be found in lab stock. If you have any questions on this post email me. If you have any comments or personal insights to add use the "Comments" section below.

Simulating High Bandwidth Power Supply Transients

A power supply transient can be defined as an unintended variation of the supply’s amplitude or current. Example terms that refer to supply transients include power surge, spikes, dropouts, interrupts, etc. Power supply transients are caused by things like a sudden sharp load change or when external energy is coupled into the supply, such as when a lightening strike occurs near the supply.

Simulating power supply transients is needed to ensure the device will work to spec in its intended operating environment and ensure it is reliable. Example applications for power supply transient testing is in automotive, aircraft, and satellite electronics. In this blog post will look at a low cost solution for simulating high bandwidth power supply transients using general purpose test equipment and some simple analog circuitry.

With modern high performance power supplies, transients with rise or fall times as fast as ~ 1ms can be simulated. The below images show a captured 10 V transient pulse on top of a 12 VDC level created with Agilent's N6705B DC Power Analyzer. You can see the measured rise time of the pulse is ~ 300 us.

To simulate power supply transients or power arbitrary waveforms with rise and fall times less than 300 us you can use the below solution, which we will call the Active Variable Resistive solution or AVR for short.

From the above AVR schematic, the two DC sources and the waveform generator can be implemented with general purpose test equipment. The MOSFET Switches and Choke are implemented using basic circuit components and should be chosen based on power levels and other considerations that we will discuss.

To go through the AVR system’s theory of operation lets use a simple example transient. Lets say we have a 10 V nominal DC power supply level powering our DUT (load) and we want to create a 10 V transient pulse with a 1 ms pulse width on top of our power supply level.

To start we would set DC source 1 to a 10 V level to serve as our nominal DC power supply level. The waveform generator would be set to a negative voltage value, like -5V, to turn on the P-type MOSFET switch so it acts as a short. In turn the negative voltage from the waveform generator would ensure the N-type MOSFET switch is reversed bias or off. Since the waveform generator’s low is floating, it is tied to the same node as the high of the load so it is at the same potential as the sources of the MOSFETs which are also connected to the same node as the load high. The choke is added to increase the impedance to ground when a transient is generated since there is a small amount of capacitance between the waveform generator’s low and ground. DC source 2 is set to 20 V, but no current is flowing out of it because the N-type switch is off. The current condition of the AVR system is DC source 1 is supplying power to the DUT at 10 V. To create the transient pulse we program the waveform generator for a single 1 ms pulse. The amplitude of the pulse should be enough to fully turn on the N-type switch, for this example we will say 5 V. When the pulse is triggered it will turn the N-type switch on and the P-type switch off. DC source 2 will quickly pull the load node up to 20 V and DC source 1 is effectively removed from the DUT. After 1 ms the waveform generator’s output will go low again and turn off the N-type switch and turn on the P-type switch. This removes DC source 2 from the load and puts DC source 1 back in the load circuit. At this point we have generated our 10 V transient pulse on top of our 10 V power supply level and we are now back at our initial conditions.

To generate a negative pulse with the AVR we would set DC source 2 for the nominal DC level and set DC source 1 to the bottom level of the interrupt, which could be any level between 0 V and the nominal DC level. Set the waveform generator so that the initial condition of the N-type switch is on and the P-type switch is off. Then, using the waveform generator, send a negative pulse to toggle the switches to create the interrupt. To create more complex power waveforms you would use the MOSFETs as variable resistors instead of switches. In this case ensure you use high power MOSFETs and heat dissipation as needed. Also for more complex waveforms a two channel waveform generator would be ideal so the two MOSFETs are not tied to the same waveform.

I implemented the solution using the following test equipment and components:
•DC source 1: N6705B DC Power Analyzer with N6762A Module
•DC source 2: N6705B DC Power Analyzer with N6762A Module
•MOSFET Switches: Si4410BDY and Si4925BDY
•Waveform Generator: 33522A Fg / Awg

Using the implemented AVR solution I created the below example 15 V 1 ms transient pulse on a 10 VDC level. The first scope shot shows the whole pulse and the second shows the rise time. Both rise and fall times were less than 6 us.

The pulse transient was generated into a 10 Ohm load with a 0.1 uF, 1 uF, and 10 uF capacitors in parallel. This was done to simulate a real world DUT supply input that presents a low impedance to ground for AC signals. Below is an interrupt created with the solution into the same load. The DC level is 25 V. The interrupt pulse is 20 V with a width of 500 us. The resulting fall and rise time of the interrupt is about 7 us.

Simulating power supply transients with a modern high performance power supply allows you to create waveforms with rise and fall times as fast as 300 us. To achieve higher bandwidths a high cost power system is typically employed by standards and quality labs. In this post we discussed and demonstrated a low cost alternative solution for creating higher bandwidth power supply transient waveforms right on the bench. If you have any questions, comments, or add-ons to this post please leave them in the "comments" section below.

Agilent Releases Seven High-Power Modules for the N6700 Modular Power System

Last week Agilent released 7 new high-power modules for the popular N6700 modular power system (optimized for system use) and the N6705B DC Power Analyzer (optimized for benchtop use). Three of the new  modules offer output power up to 300 W and the other four up to 500 W. The new modules bring the grand total of modules available for the N6700 / N6705 family up to 34, ranging from 18 to 500 W.

The new modules include advanced features such as:

  • Fast output changes (0 to 50 V in less than 2 ms)
  • Autoranging output capabilities 
  • Built-in voltage and current measurement digitizers (My personal favorite feature)
  • Optional polarity reversal relays
  • Power arbitrary waveform capabilities 
  • Active down programming capabilities
The N6705B and N6700 mainframes allow you to mix and match up to 4 modules per mainframe. The mainframes provide hardware timed output sequencing, at turn-on or turn-off, of the four power outputs. For larger channel needs, sequencing can be setup across multiple mainframes. Also the power outputs can be placed in series or parallel for increased power needs.

Below is a list of the new modules and their power, voltage, and current capabilities. For more information on each module, click on the model number to go to its product page.

Simulating Power Transients for Testing ECUs

Electronic control units (ECUs) used in automotive and aerospace/defense applications need to be immune to the harsh power systems in which they operate. Power system surges and drop-outs are common, so you need to thoroughly validate your ECU to assure proper operation. To assist ECU designers, standard ISO test specifications have been developed that replicate the power transients seen in automotive applications. These test specifications are rigorous, and the test equipment required to generate these transients is specialized and expensive. As a result, this equipment typically remains in the quality control (QC) lab, and it may not be accessible to the design engineers who need it most.

In this post we will look at how modern high performance power supplies and their arbitrary waveform (arb) generation capability provide an easy to use and capable platform for power transient testing of ECUs. To do this we will generate two example waveforms common in the automotive industry. The focus of each example will be ease of use (no code) and the power supply's arb performance. To generate the waveforms we will use the N6705B DC Power Analyzer. N6705B is a modular high performance power supply with up to 4 supply outputs and over 20 modules to choose from. Outputs can be put in series or parallel for higher voltage and current needs. For creating power arbs 50 W and above the N675xA series of modules is the best choice for ECU power transient testing. The N675xA series can generate arbs with edge rates of ~33V/ms into a full resistive or capacitive load up to 680 uF.

The first figure below shows a power supply reset test pulse train that is commonly used for ECU test. In this case, the N6752A module is used to create a simple pulse-train using a sequence of pulses (created right from the front panel). The device under test (DUT) was a load of 100 Ω in parallel with 10 μF. A close up of the final pulse shows a rise time of 553 μs (second figure below). Fall time (not shown) was measured to be 206 μs. Rise and fall times of approximately 1 ms are commonly called for in these types of tests. 

Next let’s generate a transient waveform for engine crank immunity testing using a high performance power supply like the N6705B. We will use the Starting Profile waveform in the ISO 16750-2 specification, which is pictured below.

At first glance this waveform seems complex but really it can be divided into four common waveforms, three ramps and one repeating sinewave. Using the N6705B and its built-in waveforms along with its waveform sequencing capability, we can easily build the engine crank waveform. Below are some screen shots (click on to enlarge) that show building the four waveforms and sequencing them together all from the N6705B's front panel.

And finally below we get to the resulting engine crank output waveform into a load of 100 Ω in parallel with 10 μF.

If you have to capture and recreate or just create complex custom waveforms this can be done fairly easy too. Typically you can capture a waveform on a scope or generate it using software like Matlab and then transfer it to the power supply via a CSV file using a remote connection or a USB memory stick. In some cases the power supply may have its own waveform editing software. For instance the N6705B has accompanying software (model number 14585A) that provides waveform editing as well as other measurement features. 

In the post we looked at how modern power supplies like the N6705B can be used to generate complex power transients for ECU testing. They provide an easy to use power transient testing alternative to design engineers who can't easily access expensive test setups in the quality control lab or who want to avoid expensive back and forth trips to compliance testing labs. If you have any personal incites or comments you want to add please use the comments section below.

Power Supply Rejection Ratio Testing Methods

Power supply rejection ratio (PSRR) is term typically used to describe the amount of DC power supply noise a device under test (DUT) can reject. PSRR testing is done on all kinds of electronic and semiconductor devices including ADCs, amplifiers, oscillators, etc. A basic PSRR test consists of three parts:
  1. Summing a noise signal onto a DC power supply
  2. Measuring the resulting output or functional response of the DUT. Typically done with a scope or spectrum analyzer.
  3. Using the known input parameters and resulting output to calculate the DUT's PSRR
In this blog post we will be focusing on part 1, summing a noise signal on the DUT's DC power supply. There are some helpful links at the end to address parts to 2 and 3.

The desired noise signal that needs to be summed to the power supply varies depending on the application, but it can range anywhere from an AC line ripple signal to a 20 MHz bandwidth Guassian noise signal. Today's high performance supplies often have arbitrary waveform capabilities built-in, such as Agilent's N6700B or N6705B modular power systems. These high performance power supply's are typically limited to arbitrary waveform bandwidths of 50 KHz or below. This makes them a good solution for PSRR testing  using noise signals up to the audio frequency range. To do PSRR testing above 50 KHz you can combine a power supply, a function / arbitrary waveform generator (arb), and some simply analog circuitry. In the following sections I will cover two similar methods for summing noise onto a power supply for PSRR testing. Both methods use a capacitor to couple or sum the noise signal from the arb onto the DC level from the power supply. The first method is called the inductive blocking method and an example setup implementing this method can be seen in the figure below.

In the above figure the capacitor blocks the DC output of the supply from the arb output and couples the noise signal onto the DC level. The 100 Ohm resistor between the capacitor and the arb's output is used to limit the current drawn from the arb. All arb's have very limited current output capability so the resistor is added to prevent over current error conditions. You want to chose a large capacitor value to ensure it appears as a short to the noise signal. The purpose of the inductor between the supply and the capacitor is to create a high impedance "block" between the noise signal and the low impedance output of the power supply. The magnitude of the noise signal on the DC level is dependent on the arb's amplitude setting, the arb's output impedance setting, the current limiting resistor value, and the impedance of the DUT. One thing to note when doing PSRR testing is you want to ensure the DUT's impedance is almost entirely resistive, meaning you should remove all bypass capacitors from the DUT's supply input. At the end of the post we will discuss alternatives if it is not possible to remove all parallel capacitance from the input of the DUT's power supply.

Let's run through a quick example using the test setup in the above figure. Our example DUT supply requirements are 5 V and it draws ~ 200 mA of continuous current so its impedance is ~ 25 Ohms (5V / .2 A = 25 Ohms). We want to add a 2 MHz sine wave at 177 mVrms (500 mVpp) to represent a switching power supply noise component (a high amplitude noise signal was chosen for example purposes). In our example the 33521A arb is being used. The 33521A allows you to set its output impedance to match the load impedance. If we assume the capacitor is a short and the inductor is an open to our noise signal, the load impedance that the arb sees is 125 Ohms (current limit resistor + DUT load impedance). We can now use ohms law and simple circuit theory to calculate what we need to set the amplitude of the arb to so we deliver a 177 mVrms noise signal to the DUT.

Arb amplitude setting: (125 Ohms / 25 Ohms) * 0.177 Vrms = 0.885 Vrms

For practical purposes you can use a scope at the DUT's supply input to fine tune the noise signal's amplitude level. Below is two screen shots from a scope showing the noise signal on top of the DC supply level. The first shows the full signal view and the second zooms in on the 2 MHz sine wave.

The advantage of using the inductor as a block is it acts like a short for DC but acts like an open for our noise signal. Where the inductor causes major problems is when the DUT has a dynamic current profile that varies sharply. In the case of DUT's with dynamic current profiles, the inductor can cause severe oscillations and sharp voltage transients on the power supply level. That brings us to the second method, the resistive blocking method. The setup is the same except the inductor is replaced with a resistor.This method is better suited for DUT's that have dynamic current profiles. Compared to the inductive blocking method, it will not cause oscillations and it will reduce voltage transients that occur from sharp DUT current swings. Of course adding a series resistance creates a severe unwanted voltage drop between the supply and DUT. To overcome this use the power supplies sense lines and place them after the series resistor. That way the power supply will automatically compensate for the effects of the series resistor ensuring a steady 5 V level is delivered to the DUT. Also this means we can no longer assume there is an open at the supply so that changes how we would calculate the amplitude of our noise signal delivered to the DUT. Power supplies have a parallel capacitance at their output that is typically greater than 5 uF. If we assume that supply's impedance to our noise signal is a short, we get a parallel impedance calculation between the resistor and the DUT's impedance.

Let's redo our earlier example calculation but this time we will replace our inductor with a 75 Ohm resistor. The 25 Ohm term in our earlier calculation is replaced with a parallel resistance value of 18.75 Ohms. This results in us having to increase the arb's output amplitude to 1.121 Vrms to sum a 2 MHz sine wave at 177 mVrms to the 5 V supply.

The following are some important tips and considerations for using either the inductive blocking or resistance blocking methods for PSRR testing.
  • Use either a linear supply or a switching supply with low output noise to ensure you do not add additional noise signals to your desired noise signal.
  • Use a supply with a fast transient response spec. This will help compensate for unwanted effects caused by the inductor or resistor in series with the supply's output.
  • You want to ensure the DUT's impedance is mainly resistive. If possible remove all bypass capacitors from the DUT's supply input. If this is not possible the DUT's impedance will appear extremely low to our noise signal, drastically reducing its amplitude. One way to compensate for this is to make the current limiting resistor value as low as possible to decrease its voltage drop. In cases where there is a large capacitance at the input of the DUT you may have to add a wide bandwidth voltage amplifier to the output of the arb.
Below you will find some helpful links related to this post. Also use the comments below if you have any tips or questions related to this post.

Power Supply Constant Voltage (CV) and Constant Current (CC) Modes

Young engineers and technicians often have trouble understanding constant voltage (CV) and constant current (CC) settings on a variable power supply when testing. Also they sometimes get confused why a power supply is only delivering 0.634 A to the load and not the 1 A that they set it for. The following video is made by a colleague of mine and it explains CV and CC modes on a power supply and how the load characteristics determine which mode the power supply is in. The first video explains from a theoretical point of view and the second one demonstrates it on a power supply and DC electronic load.


Making AC Impedance Measurements on Fuel Cells

Making AC impedance measurements on fuel cells can help identify problems with the fuel cell components and help identify deviations in the fuel cell assembly process. Typically when testing fuel cells, multiple impedance measurements are made at various frequencies, the results are then plotted across the frequency band resulting in an Electrochemical Impedance Spectroscopy (EIS) measurement. In this blog post we will cover building a test system for making AC impedance measurements on fuel cells

To make AC impedance measurements on a fuel cell that is producing dc current we first need stimulate the fuel cell output with a load that is varied sinusoidally at a particular frequency and then digitize the cell's output AC current and AC voltage. The digitized signal data would then by fed to a PC where custom software or a math package, such as Matlab or Excel, would be used to perform post processing on the digitized data that includes:
  1. Perform a Fast Fourier Transform on the digitize current and voltage waveform data.
  2. Using complex math functions, divide the transformed voltage by the transformed current to obtain the complex impedance (both magnitude and phase).
  3. Repeat at multiple frequencies to create a full spectrum of impedance measurements on the fuel cell.
The digitized AC current and voltage measurements needed for the impedance calculations can be done with just two pieces of equipment, a high performance electronic load (eload) and a function generation. Below shows the fuel cell impedance measurement set-up (click to enlarge).

By connecting the function generator to the eload's external analog programming input, we can use it to control the eload's load value or input impedance. By setting the function generator output to a sine wave at various frequencies we can create the necessary AC load variations to create the desired output current and voltage waveforms from the fuel cell under test. Agilent's 33210A function generator is a good cost effective fit for this application. For the eload, the reason I underline "high performance" in the above paragraph is because the eload requires some advance features including parallel voltage and current measurement digitizers built-in to capture the input voltage and current simultaniously. Otherwise we would have to add a two channel external digitizers to capture the AC voltage and current (for current we would also need a current shunt) from the fuel cell. The above figure shows Agilent's N3300 eload series. The N3300 series fills the role of a "high performance" eload because it provides simultaneous voltage and current digitization capability. The max. frequency you can make an AC impedance measurement on is going to be limited by the eload's max bandwidth specification, which is the max.rate you can change its input load or impedance setting. For the N3300 eload series the max. sinusoidal bandwidth is 10 KHz. Because of their design, eloads cannot operate at full specifications at low voltages, typically < 3 V. To over come this characteristic of eloads a "boost" power supply can be added to the circuit in series as shown in the figure. For more info on using an eload with a boost supply click here

Unfortunately going into the details of implementing an FFT in software and performing division of complex equations to make AC impedance measurements and creating a Electrochemical Impedance Spectroscopy plot is beyond the scope of this blog post. The good news is there are plenty of information on the internet for performing these calculations (just google it). Also the first link below is to an Agilent app note that provides more detailed information on preforming AC impedance measurements on fuel cells. The second link below takes you to a complete turn-key solution that provides the hardware and software needed to make AC impedance measurements on fuel cells. 


Agilent Releases New Mobile Device and Battery Test Solution

Today (June 15th 2011) Agilent release the new N6783A which is available in two versions:
  • N6783A-BAT: designed for charge and discharge testing of mobile device batteries. 
  • N6783A-MFG: designed for mobile device manufacturing to simulate the battery and characterize the mobile device’s current draw. 
The N6783A-BAT and N6783A-MFG modules are made for the N6700B Modular Power System mainframe and N6705B DC Power Analyzer mainframe. The N6700B is optimized for system test with a 1U full rack size. N6705B is optimized for bench-top use with all controls accessible from the front panel and a scope like display for analyzing digitized current and voltage measurements. Here is a closer look at the N6783A-BAT and N6783A-MFG modules.

The N6783A-BAT is a building block for handheld device battery testing:

  • Basic programmable electronic load for discharging batteries.
  • Basic programmable power supply for charging batteries.
  • Built-in digitizing measurement system.
  • Suitable for battery validation and conditioning.
  • Complimentary, low-cost alternative to the N6781A for battery cycling only.
  • For more info on the N6781A check out the June 27th 2010 post.
  • 8 V max with the ability to source 3 A of current (charge) and sink 2 A of current (discharge), see figure below

The N6783A-MFG mobile device test solution for manufacturing. This solution is equivalent to Agilent’s 66300 family of mobile comms power supplies but is in a modular form for the N6700B and N6705B platforms. Here are some of its features:
  • Excellent voltage transient response, < 75 mV of droop, < 45 µs response time in response to 0.15A / µs transients.
  • Built-in digitizer for fast, accurate, flexible measurements that are customizable to the level of speed and accuracy desired.
  • 18 W of power, voltage 6 V, and Current at -2 A, 0 to 3 A.
Since the N6783A-BAT and N6783A-MFG are modular you can mix and match them with any of the over 20 power modules available for the N6700B and N6705B mainframes.


Tools for Low Power Design of Wireless Sensors

Agilent’s N6781A and N6782A SMU modules, for the N6700 and N6705 modular power systems, provide cutting edge technology for battery drain analysis and low power design. These SMUs provide low current measurement capability, fast transient response, and digitized measurements for characterizing dynamic current. But what sets these solutions apart from anything else on the market is their seamless current ranging capability. The seamless current ranging capability allows the N6781A and N6782A to seamlessly change current measurement ranges without any discontinuities in the output. See my 6/27/10 post for more details on the patented seamless current ranging capability.

The N6781A and N6782A were specifically targeted at the mobile phone and smart device markets, but they are also proving an ideal solution for other battery powered devices such as wireless sensors. In this post I want to share some details on how a designer of wireless temperature / humidity sensors used the N6782A SMU module as a tool for optimizing their design for low power consumption. Below is a screen shot from the 14585A software using digitized measurements from the N6782A SMU module (click to enlarge).

The 14585A software, among other things, provides a scope like display of digitized current and voltage measurements from the N6705B mainframe. In the screen shot above you can see two current pulses representing the data transmit cycle of the sensor under test. The pulses peak around 14 mA with a sleep current around 4 uA. In the screen shot below we zoomed in on the start of one of the pulses. As you can see the ~400 ms pulse is really a group of pulses representing transmitted bits. Circled in red is a 2 mA step representing an LED turn-on. As you can see the LED turns on about 15 ms before the transmit cycle actually begins. By delaying the LED turn-on by ~15 ms the designer could lower the sensor’s power consumption and increase battery life.

When we used the N6782A to capture the current profile of the T/H sensor we used the seamless current ranging. When the T/H sensor was in a sleep state (current in uA region) the N6782A was using its 1 mA current measurement range as the current consumption began to shoot up because of the start of a data transmission the N6782A seamlessly switched to its 100 mA current measurement range. This means we captured 18 bits of measurement resolution throughout the current capture. Let’s use that resolution to zoom in on the sleep current as shown in the screen shot below.

The N6782A is sampling at 5 us period and the screen shot above is showing about 1.5 s worth of data. Since there are too many data points to view the 14585A software decimates the data and shows three traces: a min, max, and average trace. Notice the current anomalies circled in red. They occurred on regular intervals of about 0.9 s. In the screen shot below we take a closer look at the anomalies in the sleep current.

Here we zoomed on the sleep current anomaly and placed markers around it. The average current around the anomaly is ~6 uA while the average current during the anomaly is ~11 uA. Since the T/H sensor runs for >1 year before a battery change, eliminating the anomaly will significantly add to the device’s battery life. The whole point being that the anomaly was easy to spot with the N6782A low current measurement capability and its high speed current digitizer. And remember all the current measurement points we looked at in this post, from the pulses to the sleep current, were captured with a single data log measurement. This was made possible by the seamless current ranging capability which ensures we got 18 bits of measurement resolution from the sleep current to the transmit pulse peak current. Without the seamless ranging we would have to make multiple data log measurements of the current, each at a different measurement range, and try and superimpose the data together. The designer of the sensor plans to use the N6782A, the N6705B mainframe, and the 14585A current analysis software to help them reduce the size and increase the battery life of their sensor design.


Agilent Introduces 4-Quadrant SMU Family

The Agilent B2900A Series of Precision Source/Measure Units are compact and cost-effective bench-top Source/Measure Units (SMUs) with the capability to output and measure both voltage and current. An SMU combines the capabilities of a current source, a voltage source, a current meter and a voltage meter along with the capability to switch easily between these various functions into a single instrument. The Agilent B2900A series of SMUs provide best-in-class performance at a lower price than ever before. They have broad voltage (210 V) and current (3 ADC and 10.5 A pulsed) sourcing capability, excellent precision (minimum 10 fA/100 nV sourcing and measuring resolution ) and high measurement throughput. They also support an arbitrary waveform generation function.
The Agilent B2900A series consists of four models, the B2901A, B2902A, B2911A and B2912A, differentiated through their available features (number of digits displayed, measurement resolution, minimum timing interval, supported viewing modes, etc.) and by the number of SMU channels (one or two) they contain. I haven't personally had the chance to test drive the B2900A yet so I can not provide any personal experiences in this new product overview. I have heard good things so far and hopefully I will be able to try them out in the near future. Below is list of key features:
  • Integrated voltage/current 4-quadrant precision source and measurement capabilities for easy and accurate I/V measurement
  • Wide coverage up to 210 V, 3 A DC/10.5 A pulse
  • 10 fA/100 nV minimum measurement resolution (6½ digits)
  • 10 fA/100 nV minimum sourcing resolution (6½ digits)
  • The 4.3” front panel color display supports both graphical and numerical view modes
  • High resolution arbitrary waveform generation (AWG) and list sweep functions (10 μs minimum interval)
  • High speed digitizing capability (maximum 100000 points/s sample rate)
  • Free application software to facilitate PC -based instrument control
  • IVI-COM drivers, and SCPI supporting conventional SMU command set for basic compatibility
  • LXI class C, USB2.0, GPIB, LAN and digital I/O interface

N6781A SMU was Named One of EDN's HOT 100 Products for 2010

Agilent's N6781A SMU for battery-drain analysis was one of EDN's HOT 100 products for 2010! Congrats to the N6781A R&D team for their innovative hard work.

The N6781A has an Agilent only patented seamless ranging technology that allows its 18-bit current measurement digitizer to switch from one measurement range to another with no discontinuities in the output. I actually briefly mention the N6781A in my last blog post and I cover its features and capabilities in detail in my June 27th 2010 post. The seamless ranging capability is key for handheld device design where power optimization is critical because it allows you to fully characterize the device's dynamic current profile as it transitions through various standby and operation modes. The N6781A can simulate the device's battery or it can serve as a zero ohm shunt for battery rundown tests. The N6782A is the sister product to the N6781A and it also has seamless ranging capabilities. The N6782A is geared more towards low power optimization at the component and circuit level.

Today's Power Supplies More Than a Battery with a Knob

In the past the majority of variable power supplies used for testing were viewed as little more than a battery with knob. They were large, heavy, and had little in the way of measurement capability. The measurement capability typically consisted of an analog display or a digital display with only three digits of resolution. The power supply user often did not trust the power supplies measurement capability so the supplies voltage was set and monitored using a DMM connected across the supply's output. If they needed to measure the supplies output current a shunt with a DMM or a scope with a current probe was used.
In the last decade the test and measurement industry has really witnessed the rise of high performance supplies and application specific supplies. High performance supplies and application specific supplies on the market today come with a whole host of advanced features here a quick summary of some of them:

When it comes to power supplies smaller is better
There are two dominate power supply designs used throughout electronics world, linear and switching. In the past in testing linear supplies were dominate because of their clean output power. Switchers on the other hand had a lot of noise on their output power. The downside of linear supplies was they had a huge bulky transformer which made them large and heavy. Although linear supply designs can still be found, the test and measurement industry has really switched to switcher supplies. Why? Because switchers are smaller and advances in electronic filtering has made switcher outputs just as clean as linear outputs. For instance, Agilent's N6700 modular power supply family can deliver up to 1200 W per mainframe in a 1U full rack size (see figure below) with outstanding output noise specs.

Throw away the current shunt and current probe
Today's high performance supplies have built-in high accuracy measurement digitizers in them for capturing voltage and current in parallel with measurement resolutions up to 18 bit. The digitized points can be integrated together for increased accuracy or the points can be used to capture sharp transients which are common occurrence in today's high speed digital electronics. This beats the current measurement methods of the past. For instance you no longer have to deal with the complexity of setting up a measurement shunt with a DMM or external digitizer and you do not have deal with unwanted series resistance that results from shunts. Using a current probe you do not have to worry about adding series resistance to the circuit, but current probes are not very accurate (typically >1% error) and they typically can't measure current below 10 mA. Below is a screen shot from Agilent's N6705B showing digitized current pulses on its scope like display.

Modern power supplies are really making waves
Modern power supplies have high speed output control loops for dealing with sudden output transients. Supply designers have learned how to manipulate these fast output control loops to create high power voltage and current waveform capabilities in high performance supplies. This has given test engineers the ability to simulate engine crank profiles, to simulate a handheld battery powered device being dropped, or to simulate power line noise on a DC level with just a power supply. 

You brake it, you buy it
Prototype cellular base stations and satellite modules are very expensive. If something goes wrong with one of these expensive devices during the test process and they suck too much current from the supply it can lead to some costly damage and design delays. In the past test engineers had to build in expensive protection circuitry between the supply and the device being tested. Today's high performance supplies have made test engineer's job easier by adding a long list of safety features. As an example, Agilent's N6700 modular power supply family has a built-in watchdog timer. The the watchdog timer can be activated and set by the user to start timing after each command received by the system software. If the timer value that the user sets runs down before the next command from the system controller the power supply will shut off its outputs. This protects the device being tested in the event that the system controller or software freezes up or crashes.

There is a power supply for that app
Some of todays advanced applications have unique power requirements. To meet these unique power challenges test and measurement vendors have developed advanced application specific supplies. An example is Agilent's E4360A Solar Array Simulator. This supply is designed to be a high speed and high power current source. Its output I-V characteristics simulate the I-V output curve of a solar panel or an array of solar panels.This type of supply is used to test satellite power systems and terrestrial solar panel max power point tracking devices (for more info check out this post http://gpete-neil.blogspot.com/2010/11/simulating-photovoltaics-with-standard.html). Another example is Agilent's N6781A and N6782A. These advanced supplies were designed for battery drain analysis of handheld devices and low power optimization of handheld devices and the components that go into them. These supplies have many advanced features but the most impressive is there seamless current ranging capability. Meaning they can measure transition from one current range to another without any discontinuities in the output power. This means they can capture a sudden burst of current that goes from uAmps to Amps with 18 bits of measurement resolution throughout the whole pulse (for more info check out this post http://gpete-neil.blogspot.com/2010/06/breakthrough-dynamic-current.html).

The supply features and the application specific supplies I just covered are just examples of the wide range of features that can be found in modern power supplies. Recently power optimization has become a major factor in the design process in the electronics industry. This emphasis on power optimization will continue to drive more advanced power supply features from the test and measurement industry well into the future. One thing is for sure the high performance power supplies of today and tomorrow are really revamping the power supplies image from a battery with a knob to a sophisticated piece of instrumentation.

Power Supply Performance when using Remote Sense

In test systems the cabling path impedance between power supply and DUT can be significant. Meters of cabling combined with switch relays can equate to large voltage drop during times of high current consumption by the DUT. Its probably no secret to you that the method to compensate for this is to use the power supply's remote sensing capabilities. But what you may not know is the effect that remote sense has on the output performance of a supply. For most test applications this degradation to performance can be ignored. When it  becomes a factor is when a DUT makes large and fast load changes during operation. An example of this would be communication equipment when it goes into a transmit state.
When in remote sense mode the power supply is a feedback control system designed to control a voltage at the DUT. The figure below represents the voltage control system in block diagram form.

"Vprog" represents the desired voltage at the DUT. The summing amplifier compares the desired output voltage with the voltage at the DUT as measured by the sense amplifier. Any difference is amplified and applied as a correction signal to the power control circuit block. This block adjusts the power supply output until the measured output voltage equals the desired output, compensating for any external voltage drop. As shown in the block diagram the load and lead impedance become part of the voltage control system.
This system provides great control of the average DC voltage at the DUT. However, the accuracy to which the instantaneous voltage can be controlled for sharp load changes depends on several factors, namely:
1. The resistance and inductance of the cable connecting the power supply and DUT.
2. The input impedance of the DUT.
3. The amplitude, rise and fall times of the load change or current pulse.
4. The bandwidth of the power supply control loop.
5. The voltage slew rate and transient response capability of the power supply.
The final result is that while the average voltage may be ideal, the instantaneous value may be
less than ideal. If the instantaneous voltage drops too low it can cause your DUT to go into a reset mode which ruins your test. There are three common ways to compensate for this: minimize path impedance, add filtering to the DUT, and use high performance supplies.
To lower path impedance (both R and L) shorten wire lengths when possible, use larger gauge wire, make good connections, use twisted pair, pay attention to contact resistance, and try to avoid hot switching. A second solution is to place a large electrolytic capacitor across the terminals of the DUT right at the test fixture.When the load goes through a sharp change the capacitor will charge or discharge to maintain a constant voltage giving the power supply time to catch up. This method does have drawbacks though such as longer settling times when a voltage change is made and distorted current measurements during sharp load transients due to the charging and discharging of the cap. Finally the best way is to just buy high performance power supplies with high control loop bandwidth and fast transient response (to see a post on the transient response spec click here). Look for transient response specs < 100 us.

Agilent Introduces 4-Quadrant General Purpose SMU

Today Agilent introduced a 4-Quadrant General Purpose SMU known as the N6784A. The N6784A is a plug-in module for Agilent's popular N6705B DC Power Analyzer and N6700B Modular Power System. The N6705B mainframe is optimized for bench-top use and the N6700B mainframe is optimized for system use. It is the first 4-quadrant module for these platforms. Features and capabilities include:

  • Two output ranges: +/-20V +/-1A or +/-6V +/-3A
  • Glitch-free operation – change sourcing ranges or measurement ranges without any glitches
  • Four current programming ranges – precisely source current down to μA
  • Stable operation with capacitive loads up to 150 μF
  • High-speed output can slew at 10 V per μs into a resistive load
  • Fast modulation of DC output – create arbitrary waveforms up to 100 kHz (sine) into a resistive load
  • High-speed digitized measurements – capture/view the power consumption of the DUT up to every 5 μs with built-in 200 kHz digitizer
I am a big fan of its emulation modes which improve usability by instantly configuring the SMU for the most common use cases. When one of the emulation modes is selected, the SMU optimizes all of its features and settings for that particular use case. Emulation modes:

  • 4-quadrant power supply
  • 2-quadrant power supply
  • Unipolar power supply (i.e. 1-quadrant)
  • CC load
  • CV load
  • Voltage measure (i.e. voltmeter mode)
  • Current measure (i.e. ammeter mode)

Sequencing Multiple Power Supply Outputs

Today’s high performance power supplies continue to add more and more capabilities to make the test engineer’s job easier. The capability that I am going to talk about here is sequencing on or off multiple power outputs. This capability allows you to set the order and timing that each power supply output powers on or off. This is useful for designing and testing embedded system designs. Embedded systems can be made up of any combination of microcontrollers, FPGAs, ASICs and memory chips. These individual integrated circuits often have multiple power input requirements that must be properly sequenced on and off to prevent latch-up. Latch-up may cause a wasteful initial surge of current at turn on, or it may be severe enough to inflict permanent damage to the semiconductor device. Ultimately these devices will have a power distribution system with regulators that will ensure the proper sequencing and timing for each power supply turn on and turn off, but during initial design and testing the power distribution system is often not in place yet so test and measurement equipment is used in its place to simulate the proper turn on and off conditions of the design.
In the past power sequencing was typically done in one of two ways: using programmable power supplies with software or using supplies with custom switches. The first way uses programmable supplies and then in software the supplies are properly sequenced on or off. The drawback to this method was unless you were running a real time operating system (windows is not a real time operating system) there was no way to accurately guarantee the timing from one supply output to the next with better than 30 ms of precision. The other method required additional hardware in the form of switch cards and control circuits. A hardware timed control circuit would provide the precision timing that a computer operating system could not. The control circuit would then control the sequencing of the supply output on or off using switches. If you wanted to avoid switch bounce you had to use solid state or mercury switches versus traditional mechanical switches. The problem with this method is complexity it adds to the testing process.
With the capability built into the supply you avoid the complexity of dealing with multiple pieces of hardware and since the timing is done inside the hardware of the supply it is highly accurate. Agilent’s N6700B series of supplies and the N6705B DC Power Analyzer are examples of supplies with built-in power output on / off sequencing. Each one has up to four power supply outputs per mainframe that can be sequenced on or off. Sequencing from one mainframe to another is also possible for applications that require sequencing of more than four power supply outputs. The figure shows a picture of the N6705B DC Power Analyzer being used to test an embedded design that requires turn on power sequencing.

You Can Get a Watchdog with Your Power Supply?

Watchdog timers are common place in the world of computing and embedded systems. It’s the watchdog timer’s job to notice if a computing device, such as a microcontroller, has hung up or froze. If it has the watchdog timer performs a reset. The whole thing works by setting the watchdog timer for a specific count. Once the computing device starts operating the watchdog timer starts its counting. It is the computing device’s job to send a command to the watchdog timer to let it know it is still running (referred to as “kicking the dog” or “feeding the dog”). Once the watchdog gets that command it resets  and starts counting again.
Recently Agilent added watchdog timer functionality to its N6700 modular power system family of supplies. The N6700 watchdog functionality works similar to the description above with the “feed the dog” timer reset coming from the IO connection with the computer. A timer reset occurs whenever SCPI traffic from the computer occurs. If the watchdog timer does not receive a reset in the user’s specified count it goes into a protection mode and shuts its outputs off.
The whole idea of adding this functionality developed from customer feedback. The customer was running durability tests on multiple DUTs. The tests lasted for weeks or months at a time. During the tests it wasn’t uncommon for the computer to freeze or crash (damn you windows). During testing the supply outputs were cycled high and low. If the computer goes down while the supplies are on the high end and there was no one around to notice the DUTs would be destroyed. From there the tests had to be started from scratch with new DUTs, very costly. The moral of the story here is do not be afraid to tell your test and measurement company reps what capabilities you need because they do listen!

Photovoltaic I-V Curve Characterization using a DC Electronic Load

This is a follow-up to my post on 6/11/10 that covered performing MPPT using a DC electronic load (eload). MPPT with an eload is done for design verification and durability testing of Photovoltaic (PV) devices like PV panels and concentrated PV. Besides just performing MPPT related testing, eloads make great solutions for I-V curve characterization for manufacturing and R&D test. They are great for testing high current PV devices like large area cells, modules, panels, and concentrated PV since eloads can sink and measure high current for a low cost. Below is a link to a video on youtube starring yours truly that covers using an eload for characterizing I-V curves.

Video link: Testing High Power Solar Cells and Modules Using Agilent's Electronic Load

Breakthrough Dynamic Current Measurement Technology for Low Power Optimization in Portable Electronics

On June 21st Agilent released two new 2-quadrant SMU modules, the N6781A and N6782A. The N6781A is optimized for battery drain analysis and the N6782A is optimized for functional test. These modules go into Agilent’s popular N6705A/B DC Power Analyzer and N6700 Modular Power System. These SMU modules provide current measurement capability down to 10s of nano amps, four different current measurement ranges, fast transient response, I and V arbitrary waveform capability, and two 200 KSa/s 18 bit measurement digitizers for capturing voltage and current in parallel. Now those are awesome features right there, but what really makes these SMUs special is a patent Agilent only technology that gives them the ability to seamlessly transition through current measurement ranges without any discontinuities or glitches in the output power being delivered to the DUT. This innovative feature is intended for engineers testing devices where low power optimization is critical for delivering maximum battery life such as cell phones, handheld radios, or portable medical devices.
The seamless measurement ranging ability means the N6781A and the N6782A can capture dynamic current ranging from milli or micro amps to amps with high accuracy and 18 bits of resolution over the whole range. For those out there who currently use current shunts to capture dynamic current this means no more worrying about shunt loading effects when the current goes to the high end of the range and no more worrying about poor accuracy and resolution when the current dips to the low end of the range. For those currently using high performance supplies to capture dynamic current this means no more running tests multiple times at different measurement ranges to get each piece of the current profile puzzle and then having to piece it together. The figure gives a visual picture of what the seamless ranging can do.

For more information on the new N6781A click here
For more information on the new N6782A click here

What is the Load Transient Recovery Specification Telling Me?

Whenever a load change occurs on the output of a power supply, that power supplies output voltage will momentarily change from its programmed value. The figure to the right is a screen shot from an scope showing the change of a supplies voltage level after its load went from 1 mA to 500 mA in about 30 us.

The specification for this power supply characteristic is called the Load Transient Recovery Time or Transient Response Time. It represents how long it takes a power supply to return to its set voltage level after a sudden change in load current.
The specification typically has three parts to it:
•Magnitude of the load change, such as from 50% of full load to 100% of full load. So if we had a power supply that was rated for a max current of 10 A the spec would be referring to a load change from 5 A to 10As
•Voltage settling band is how close the voltage level will settle to its original level before the load change. Note that after a load change the power supply’s never recovers to its original level. How close it gets to returning to its original level is dependent on the magnitude of the load change.
•The time it takes the power supplies voltage level to settle within the voltage settling band

Below is an example of a transient response specification for two of Agilent’s N6700B modular power system’s supplies.

N6751A & N6761A Transient Response
Magnitude of load change: 60% to 100% and from 100% to 60% of full load for models N6751A & N6761A
Voltage settling band: ± 75 mV
Time: < 100 μs

The example spec can be interpreted as the output of the N6751A will return to no more than 75 mV within its original value in less than 100 us when a load transient that is 60% to 100% or 100% to 60% of its full scale load current occurs.

The transient response time is highly dependent on the speed of the supplies internal voltage output monitoring loop. Speeding up this loop provides better transient response time, but the output becomes more susceptible to instability and oscillations. That is why supplies with short transient response times typically cost more because of the extra investment in engineering it to ensure good output stability.

If the load is changing too fast for the supplies transient response time to keep up the power supply will never be at its programmed voltage level. With that in mind, the transient response time of a power supply is strongly related to the output bandwidth of the supply. The reason you typically do not see an output bandwidth spec is because the bandwidth of the supply is dependent on its load and power supply manufacturers cannot predict what type of load their customers will connect to the output of their supplies.

Eloads are Rreat for Outdoor Photovoltaic Test but Where is the MPPT Capability?

Eloads have become a popular solution for outdoor testing of higher power photovoltaic (PV) devices, like PV panels and concentrated PV. The main reason for this is eloads can sink a lot of current at a low cost compared to 2 and 4 quadrant power supplies. The testing is usually of the design verification variety and one of the main roles of the eload is max power point tracking (MPPT) on the output of the PV device. One request of end users of eloads for this application is does the eload have MPPT capability or can you put built-in MPPT capability in the eload? Currently there are no general purpose eloads (that I know of) that have built-in MPPT capabilities. This means it is up to the test engineer to implement an MPPT algorithm in software. This adds time and complexity to the test engineer’s job. Also since the algorithm is in the software it has to deal with IO latency between the computer and the eload which lowers the test systems MPPT speed. To help test engineers out with this challenge I wrote an article entitled “A Photovoltaic MPPT Algorithm for DC Electronic Loads” that was published by Electronic Design and can be found at the link below. The article introduces an ideal algorithm for performing MPPT with an eload. It is ideal because of its low complexity and it keeps IO transactions to a minimum to reduce the affects of IO latency.

A Photovoltaic MPPT Algorithm for DC Electronic Loads